Joint optimisation of supply and bias modulation

ABSTRACT

There is disclosed a technique for controlling at least one amplification stage, comprising: selecting a linearity objective for the amplification stage; in dependence on an input signal to said amplification stage, determining a combination of supply input and bias input for the amplification stage in order to meet said linearity objective; and in dependence on there being more than one combination of supply input and bias input for meeting the linearity objective, selecting the combination that optimises a further system performance objective for the amplification stage. The further system performance objective may be one or more of: an efficiency objective; an envelope signal bandwidth objective; or a robustness to production tolerance objective.

CROSS REFERENCE TO RELATED APPLICATIONS

This case is a continuation-in-part of pending U.S. patent applicationSer. No. 12/293,315 (Attorney Docket: 9811-004US), filed Sep. 17, 2008,which is a 371 of international PCT/GB/2007/000959, filed on Mar. 19,2007, which claims priority of GB Application 0605472.0, filed on Mar.17, 2006, each of which is incorporated by reference herein.

FIELD OF THE INVENTION

The invention relates to techniques for achieving amplification ofnon-constant envelope signals. The invention particularly, but notexclusively, relates to the amplification of radio frequency (RF)signals.

DESCRIPTION OF THE RELATED ART

Many modern communication systems typically use non-constant envelopemodulation techniques to achieve high spectral efficiency. To avoidspectral spreading into adjacent communication channels, high linearityradio frequency (RF) amplification is required. Traditional fixed biasamplifiers can only achieve the required linearity by ‘backing off’ theamplifier so that it normally operates at a power well below its peakpower capability. Unfortunately, the DC to RF power conversionefficiency in this region is very low. As a consequence these designsdissipate considerable heat and reduce battery life when used inportable applications.

Maximisation of battery life is of paramount importance in mobilewireless equipment. With most high spectral efficiency communicationstandards, the mobile transmitter operates at considerably less thanmaximum power most of the time. There are two reasons for this. Firstly,power control is generally used to reduce the average transmit power tothe minimum level required for reliable communication, and secondly,most emerging modulation schemes have a high peak-to-average powerratio. Hence it is important for the power amplifier to maintain highefficiency at powers significantly below maximum, where the poweramplifier operates most of the time.

A well known prior art technique for increasing amplifier efficiency,known as “envelope tracking” (ET), uses a supply modulator to modulatethe supply voltage substantially in line with the envelope of the inputRF signal (Raab F. H., “Efficiency of envelope tracking RF poweramplifier systems” Proc. of RF Expo East, Boston, USA November 1986, pp.303-311). Classically, a voltage margin is added to the dynamic supplyvoltage to ensure that the power amplifier always operates in linearmode. To achieve highest overall efficiency, the efficiency of thesupply modulator itself must be high, requiring the use of a switchedmode DC-DC converter for the modulator. The design of the supplymodulator is critical to the system performance of the amplifier. Inaddition to achieving good efficiency, the modulator must also exhibithigh bandwidth, high linearity and low noise to be useful in moderncommunications applications which typically use high bandwidth CDMA orOFDM modulation schemes and also demand high modulation accuracy.

One prior art technique for the supply modulator design (commonlyreferred to as a class-S arrangement) uses switch mode pulse widthmodulation (U.S. Pat. No. 6,141,541, U.S. Pat. No. 6,025,754). Althoughpractical for low modulation bandwidths, in such class-S arrangementsswitching losses become unacceptable at the rates required for modernmodulation formats.

Another prior art technique for a supply modulator design (commonlyreferred to as a class-G arrangement) uses multiple voltage sources anddynamically switches the amplifier supply terminal between the sourcesdependant on the instantaneous envelope level (WO 0118956, U.S. Pat. No.5,115,203). However, a drawback is that the instantaneous switchingcreates noise and intermodulation distortion (IMD) products in the RFoutput which are difficult to remove. A modification to this techniqueuses linear interpolation between the switching levels to greatly reducethe noise and IMD products (WO 2004/075398).

Another well known prior art technique for increasing amplifierefficiency is to dynamically modulate the RF amplifier biassubstantially in line with the envelope of the modulating signal (U.S.Pat. No. 4,462,004). Although some improvement in efficiency can beobtained by using dynamic biasing, this is significantly less than canbe achieved by supply modulation. It has also been proposed that dynamicamplifier bias modulation may be used in conjunction with supplymodulation to improve efficiency (WO 03056698).

Other techniques can also be used to improve efficiency. EnvelopeElimination and Restoration (EER) uses a limiter to remove all amplitudemodulation (AM) on the RF input signal, and then re-applies the AM usingsupply modulation of the RF amplifier (WO 9905783). This techniqueoffers good power added efficiency (PAE) improvement at high signallevels, but relatively poor PAE at lower signal levels due to a highinput drive level. It also suffers from several significantimplementation problems including capacitive leakage from input tooutput at low signal levels, which degrades modulation accuracy, and theneed for the supply modulator bandwidth to be significantly greater thanthe envelope bandwidth.

Alternatively, the amplifier device periphery can be altered to improveefficiency (U.S. Pat. No. 6,445,247). Although such a technique may beeffective as a means of tracking slowly varying changes in averagepower, it is less effective as a means of enhancing efficiency withsignals having high Peak-to-Average Power (PAP), such as OFDM signals.This is because of the problem of achieving smooth transitions in deviceperiphery, without which noise and IMD targets is difficult to solve.

In summary, from the known prior art arrangements, ET shows promise as aviable efficiency enhancement solution. However, a disadvantageous sideeffect of supply modulation is that if the supply voltage preciselytracks the envelope, or is optimised for best amplifier efficiency ateach envelope level, the RF gain reduces at low input levels. Thenon-linearity so introduced results in the generation of unwanted IMDproducts. Various techniques have been proposed in the prior art toameliorate these effects. These include pre-distortion of the RF input(WO 02058249), and the use of envelope feedback from the RF output (US2003/0045238).

An alternative linearization approach is to use an envelope voltage tosupply a voltage mapping function to achieve constant gain from the RFamplifier, thereby reducing the need for pre-distortion or feedback (WO0118956). The mapping function between envelope voltage and supplyvoltage may use a continuous function, in which the envelope voltage maybe uniquely derived from knowledge of the supply voltage, or usethresholding, whereby the supply voltage is held constant when theenvelope falls below a prescribed level (U.S. Pat. No. 6,437,641).

Combinations of techniques may also be used. Dual bias (supply andgate/base) modulation schemes are described in WO 0118956, WO 0041296and “High Efficiency Class-A Power Amplifiers with a Dual-Bias-ControlScheme”, Kyounghoon Yang, George Haddad and Jack East, IEEE Transactionson Microwave Theory and Techniques, Vol. 47, No. 8, August 1999. Thesetechniques offer efficiency improvements over the use of supply or biasmodulation alone. The solution shown in WO 0041296 describes the use ofdual bias in conjunction with pre-distortion linearization and feedback.The use of a pure class-G supply modulator in conjunction with biasmodulation to achieve constant gain from an RF amplifier is described inWO 0118956. This solution does not address the noise and IMD problemsintroduced by the stepped supply voltage.

Although dual bias modulation offers attractive potential performanceimprovements, the scheme used to control the supply and bias voltages iscritical to its success.

An RF amplifier may in general be considered as a ‘black box’ systemwith a number of input ports and a number of output ports. Usually theamplifier has one RF input port, one RF output port, and two bias inputports; the gate/base bias input port and the supply bias input port.From the discussion of the prior art it will be clear that two commonsystem design objectives are to achieve high PAE and high amplifierlinearity.

The aim of the invention is to provide a method and apparatus forcontrolling an amplifier to achieve prescribed performance objectives.

SUMMARY OF THE INVENTION

The invention provides for the derivation of dual control voltages tooptimise amplifier system performance. Without the solution provided bythe invention, the complex and interdependent nature of key amplifierperformance parameters (particularly gain, phase, efficiency) withrespect to both supply and bias inputs, limit the usefulness of a dualbias architecture.

It should be noted that for the purposes of the definition of theinvention the terms supply and bias inputs are used, and for thepurposes of the description of preferred arrangements of the inventionthe terms supply and bias voltages are used. In general a supply inputmay be a supply current or a supply voltage, and a bias input may be abias current or a bias voltage. The described embodiments herein are inthe context of supply and bias voltages.

In accordance with one aspect of the invention there is provided amethod of controlling at least one amplification stage, comprising:selecting a specific system performance objective; and in dependence onan input signal to said amplification stage, selecting a supply inputand a bias input for the amplification stage in order to meet saidobjective.

Preferably one of the supply input and the bias input is optimised, andthe other is maximised.

The system performance objective is dual, meeting linearity andefficiency. In a preferred arrangement, a predefined request forlinearity is met, and then subjected to a best efficiency. A bestefficiency is thus achieved for a specified linearity. Preferably acertain value of linearity must be achieved, and then at least a certainobjective of efficiency is achieved, and preferably maximised. Boththese goals can be achieved with a joint optimisation of supply and biasin accordance with the invention.

The techniques described open up the possibility of using an amplifierto ‘self linearise’, thereby reducing or eliminating the need forpre-distortion. This is particularly attractive for mobile equipment,where increased complexity frequently carries a cost or powerconsumption penalty.

Low power RF amplifiers may be well described by a quasi-static,memory-less model described by the AM-AM (AM=amplitude modulation) andAM-PM (PM=phase modulation) performance of the amplifier. This istypically a complex function of a large number of amplifier parametersincluding device technology, device periphery, temperature, gate/basebias, supply voltage, input power and load impedance. Using automatedmeasurement techniques it is possible to build a comprehensive map forthe device of AM-AM, AM-PM and PAE performance with respect to key inputparameters, including but not limited to gate/base bias, supply bias andinput power. It is then possible to search a measurement database todetermine optimum loci for gate/base bias and supply voltage to meetspecific system performance goals. Hence the mapping function betweeninput envelope and supply voltage, and between input envelope and biasvoltage, to meet specific performance goals may be uniquely determined.

As an example, it may be desired to determine the optimum supply voltageand bias voltage locus to achieve best PAE for a wide range of outputpowers. Alternatively, it may be desired to determine the supply andbias loci giving best PAE subject to achieving a constant target gainover a wide range of output powers. Many other system performancetargets could be specified, including best PAE subject to achievingconstant phase with respect to output power.

It is also possible to formulate more sophisticated linearity targetsinvolving both amplitude and phase and to combine these with efficiencyconstraints. Minimisation of ACPR is one such example and can bedirectly calculated from instantaneous measured AM-AM and AM-PMcharacteristics.

The described techniques for determining bias and supply voltage locimay also be used in conjunction with a variety of known feedback andfeed-forward techniques to improve performance with respect totemperature fluctuations and unit-to-unit variations.

Software controlled automated equipment may be used to performmeasurements on said amplifier. A software program may be used to assistsearching of said measurement database or exploration of said model. Theoptimum bias voltage and the optimum supply voltage with respect toinput power may be approximated by separate non-linear mappingfunctions. Said non-linear mapping functions may be updated inaccordance with the temperature of said RF amplifier.

Pre-distortion of the RF waveform may be used to further optimise thesystem performance objectives of said RF amplifier.

Feedback from the output of said RF amplifier may be used to assistupdating of said non-linear mapping functions. The non-linear supplymapping means and said non-linear bias mapping means may be updated inaccordance with the temperature of said RF amplification stage.Pre-distortion of the RF waveform may be used to further optimise thesystem performance objectives of the RF amplification stage. Feedbackfrom the output of said RF amplifier may be used to assist updating ofthe non-linear supply mapping means and the non-linear bias mappingmeans.

BRIEF DESCRIPTION OF THE FIGURES

The description is accompanied by the following drawings:

FIG. 1 is a block diagram of an arrangement for providing jointoptimisation of supply and bias modulation in accordance withembodiments of the invention;

FIGS. 2 to 4 are examples of parameters measured during characterisationof an amplification stage in accordance with embodiments of theinvention;

FIG. 5 is an example of output data extracted from a measurementdatabase based on the characterisation of an amplification stage inaccordance with embodiments of the invention;

FIGS. 6 and 7 represent the measured performance of an exampleamplification device controlled to operate in accordance withembodiments of the invention;

FIG. 8 illustrates a definition of instantaneous error vector magnitude;

FIG. 9 summarises in tabular form the results of FIG. 7;

FIGS. 10 and 11 illustrate measured compression characteristics of anamplification stage adapted to operate in accordance with embodiments ofthe invention;

FIGS. 12 and 13 show predicted constellation and predicted spectrumrespectively of an amplification stage modified to operate in accordancewith an embodiment of the invention;

FIGS. 14 to 16 show exemplary implementations in accordance withembodiments of the invention.

DESCRIPTION OF THE PREFERRED EMBODIMENTS

The invention is described herein by way of reference to particularpreferred embodiments. The invention, and embodiments thereof, isparticularly advantageously suited to cost, space and power constrainedmobile applications, but is not exclusively applicable thereto.

FIG. 1 is a block diagram of an amplifier system embodying the conceptsof the invention. Referring to FIG. 1, there is illustrated anamplification stage 100, an envelope detector stage 103, and a voltageselection stage 101. The amplification stage receives an RF signal to beamplified at a first input port thereof on line 141, a supply voltage ata second input port thereof on line 108, and a bias voltage at a thirdinput port thereof on line 110. The amplification stage 100 generates anamplified RF output signal at an output port on line 142.

The envelope detector 103 receives the RF signal to be amplified on line141 at its input, and generates a signal representing the envelope ofthe RF input signal to be amplified at its output on line 109. Theenvelope signal on line 109 is then provided as an input to the voltageselection stage 101. In accordance with the principles of the presentinvention, as discussed in detail hereinbelow, the voltage selectionstage 101 generates the supply voltage and bias voltage at outputsthereof on lines 108 and 110, for applying to the second and third inputports of the amplifier stage 100, in dependence on the envelope signalon line 109. As will be described in detail hereinbelow, the voltageselection stage supplies and modulates the supply and bias voltages forthe amplification stage such that they are jointly optimised to meetprescribed linearity and performance goals or objectives. In general, inaccordance with embodiments of the invention once a linearity objectiveis met, a further performance objective of the system is then optimisedwithout losing the achieved linearity objective.

Examples of linearity objectives include spectral purity, constant gain,constant phase, minimum modulation error vector magnitude or anycombination of these. Examples of performance objectives includeefficiency objectives, envelope signal bandwidth objectives, andobjectives for robustness to production tolerance, or any combination ofthese.

In the following discussion of exemplary arrangements, the performanceobjective is discussed with reference to the example of optimisingefficiency. However this is just an example, and as mentioned above theperformance objective may be defined in dependence on other criteria.

The amplification stage 100 may be a single stage or multiple stageamplifier design with one or more supply voltage ports and one or morebias voltage ports, as required. In general the amplifier stage may beimplemented using either one of, or a combination of, device familytypes (e.g. bipolar, FET, etc.) and implementation technologies (e.g.Si, GaAs, etc.). The bias voltage may be a base bias voltage for bipolarimplementations and a gate bias voltage for FET implementations.

In accordance with the present invention, the voltage selection stage101 provides the bias voltage and supply voltage to the amplifier stage100 in dependence on the envelope of the RF signal to be amplified in away such that the efficiency of the amplification stage 100 is improved.

By characterising the amplification stage the performance of theamplification stage for different supply voltage levels and bias voltagelevels for a given input (envelope) signal level is determined. In thisway it can be determined the optimum supply voltage and bias voltagelevels for a given input (envelope) signal level.

In a preferred embodiment the optimum loci for bias and supply voltagesto meet specific system performance goals are determined by acomprehensive characterisation of the performance of the amplificationstage 100. This preferably requires multiple power sweepcharacterisations of the amplification stage 100 for variouscombinations of supply and bias voltage. The characterisation may bemade using either continuous wave or modulated RF carrier(s), and mayinclude measurement of several parameters from the followingnon-exhaustive list: supply voltage, bias voltage, gain, phase, current,input power, output power, adjacent channel power (ACP), error vectormagnitude (EVM) and correlation coefficient (p). These parameters aredependent variables, i.e. variables which are dependent upon theamplification stage.

In general, the parameters of the device are measured which arenecessary to determine a particular performance characteristic. If, forexample, it is desired only to ensure that the gain of the amplificationstage is optimised, then only those parameters necessary to determinegain are measured, for different input (envelope) signal, bias voltageand supply voltage combinations.

By way of example, particular details of exemplary measurements for asample amplification stage, and the utilisation of such measurements toachieve the aims of the embodiments of the present invention, arediscussed below. For the purposes of these examples it is assumed thatthe amplification stage comprises a single stage 1W heterostructurefield effect transistor (HFET).

The characterisation at the amplification stage 100 may be carried outin a variety of ways. The purpose of the characterisation process is toobtain measurements for the amplification stage 100 which are necessaryto determine a particular performance characteristic for combinations ofinput (envelope) signal level, bias voltage, and supply voltage. Mostconveniently the characterisation may be carried out using automatedtest equipment. As an example, the automated measurement of varioussimple parameters of the example single stage 1W HFET amplifier may beexpressed by the following pseudo-code:

For Vcc = 1 to 8V in 50mV steps  For Vbias = 1.1 to 1.9V in 10mV steps  For Pin = −5 to 21 dBm in 0.2dB steps  Record Vcc, Vbias, Pin, Gain, Current, Phase   end  end end

This ‘quasi-static’ characterisation clearly does not account for memoryeffects. Memory effects are the effects caused by a previous value of ameasurement point, i.e. the effect of a transition from a previous valueto a current value. However memory effects are of much less significancein low power arrangements than in high power devices, and thus in lowpower arrangements quasi-static characterisation is satisfactory. Wherenecessary, the characterisation process may be extended to account formemory effect.

Automated characterisation may also easily be extended to include otherindependent variables, i.e. variables independent of the amplificationstage 100 itself, such as temperature and frequency. In general it canbe understood that ‘dependent’ variables refers to those variables whichare directly dependent upon the amplification stage 100 itself, whilst‘independent’ variable refers to variables which are not dependent uponthe device but rather are dependent on features external to it.

A measurement database for a given amplification stage may thus beestablished following a characterisation process. The resultingmeasurement database can be queried to determine key aspects of deviceperformance. Parameters derived from the measured parameters, such asPower Added Efficiency (PAE) which is the difference between RF outputand RF input power divided by DC power, may also be conveniently addedto the database.

Simple non-exhaustive examples of obtained measurements for the examplesingle stage 1W HFET amplification stage are shown in FIGS. 2 to 4.

FIG. 2 shows a three dimensional plot for the result of an automatedmeasurement of the amplifier phase variation with respect to input power(Pin) and supply voltage (Vcc) for one particular fixed bias voltage(1.5V).

FIG. 3 shows a three dimensional plot for the result of an automatedmeasurement of the amplifier gain variation with respect to input power(Pin) and supply voltage (Vcc) for a particular fixed bias voltage(1.5V).

FIG. 4 shows a three dimensional plot for the result of automatedmeasurement of the amplifier power added efficiency (PAE) variation withrespect to input power (Pin) and supply voltage (Vcc) for a particularfixed bias voltage (1.5V).

Each of the three-dimensional surfaces shown in FIGS. 2 to 4 representsone of many such amplifier phase variation, amplifier gain variation andPAE surfaces respectively parameterised by bias voltage. Thus one suchsurface will result for each distinct bias voltage for whichmeasurements are taken. Where the pseudo code stated above is used,measurements are taken for bias voltages between 1.1V and 1.5V in 10 mVsteps.

From the example measurements obtained, as illustrated in FIGS. 2 to 4,for each discrete value of bias voltage there is obtained a sequence ofresults showing the variation of phase, gain and PAE with respect tosupply voltage and input power (which is representative of the inputenvelope).

As a result of such measurements, a measurement database can beestablished based on all the results obtained.

FIG. 5 illustrates an example of the use of values contained in themeasurements database. FIG. 5 shows the loci 402 for supply voltage andthe loci 400 for bias voltage versus input envelope for an amplifieroptimised to achieve maximum efficiency for a specified constant gaintarget. The loci are formed based on the discrete measurements which areobtained from the measurements database and plotted. In the example ofFIG. 5 the target gain is set as 10 dB. Thus it can be seen from FIG. 5that for a given instantaneous input envelope value, the supply voltageand bias voltage can both be selected to optimise maximum efficiency fora specified gain target. Thus the measurement database created from theinformation in FIGS. 2 to 4 may be queried to determine the combinationof supply and bias voltages which maximise the amplifier efficiency forany given input power.

FIG. 6 shows a plot of the measured gain against output powerperformance of the example single stage 1W HFET amplifier optimised fora) maximum PAE as illustrated by curve 304 and b) constant gain asillustrated by curve 302, compared with (c) a conventional (prior art)fixed bias amplifier as illustrated by curve 300. This is notillustrative of the use of the data from the measurements database, butrather illustrative of the performance achieved by the amplificationstage in use when the supply and bias voltages are selected according tothe principles of embodiments of the invention.

The measurement database may also be queried to determine thecombination of supply and bias voltages which maximise the amplifier PAEfor any given input power, using results obtained in thecharacterisation process exhibited by FIG. 4.

Curve 304 in FIG. 6 thus shows the amplifier gain associated withoptimum efficiency operation. It can be seen that at low output powerthe gain reduces substantially. This introduces two significantproblems. Firstly, the gain expansion caused by the reduced gainintroduces significant distortion, and secondly the overall efficiencyof a multi-stage amplifier is reduced as the contribution of driver andpre-driver stages to efficiency becomes more significant. One approachto dealing with this problem is to use pre-distortion linearization tocompensate the gain of the amplifier. While this addresses thedistortion problem, the efficiency problem remains and may be furthercompounded by the additional power consumption of a linearizer.

Another approach to addressing this gain expansion problem is toformulate a measurement database query to determine the combination ofsupply and bias voltages which result in a particular target gain. Thereare a large number of combinations of supply and bias voltage whichfulfill this criterion; so for example it may be further stipulated thatthe combination of interest is that which achieves maximum PAE. Such anexample loci is shown in FIG. 5. This then uniquely identifies the lociof supply and bias voltages with respect to input power.

There are many queries which could be constructed to achieve varioussystem objectives. For example, a query could be constructed whichdetermined the supply and bias voltage loci required to achieve max PAEfor constant phase.

The queries may be constructed using many software tools, such asMatlab. An implementation using SQL (Structured Query Language) for theexample HFET amplifier is now described below by way of example. Thisexample relates to a specific example query.

First, a query is constructed to create a table of measurements whichresult in 10 dB±0.5 dB amplifier gain. This query may be structured asfollows:

SELECT   [Measdata].Pband,   [Measdata].Pin,   [Measdata].Vcc,  [Measdata].Vbias,   [Measdata].PAE,   [Measdata].Gain,  [Measdata].Phase INTO   [10dBgain] FROM   [Measdata] WHERE Round([Measdata].Gain,0) = 10 ORDER BY   [Measdata].Pband;

Next, the measurement which results in max PAE for each output powerlevel is isolated:

SELECT   [10dBgain].Pband,   [10dBgain].Pin,   [10dBgain].Vcc,  [10dBgain].Vbias,   [10dBgain].PAE,   [10dBgain].Gain,  [10dBgain].Phase,   [10dBgain].ID  INTO   [constgainlocus]  FROM  [10dBgain]  WHERE   ((([10dBgain].Pband & “ ” & [10dBgain].PAE)  IN (SELECT   [10dBgain].Pband & “ ” & Max([10dBgain].PAE)  FROM  [10dBgain]  GROUP BY   [10dBgain].Pband)))  ORDER BY  [10dBgain].Pband;

This query, in effect, picks the best linearity, and then picks the bestefficiency. Linearity may be spectral linearity, constant gain (or lowvariation gain that meets linearity constraints), constant phase (or lowvariation phase that meets linearity constraints), or a combination ofany of these. Efficiency is one example of a performance characteristic.

Referring now once again to FIG. 5, which shows the resulting supplylocus 402 and bias voltage locus 400 plotted against the envelope of theinput voltage, it can be seen that the required loci may be approximatedwell by a 3^(rd) order polynomial.

FIG. 7 shows the measured efficiency versus output power performance ofa controlled dual bias amplifier optimized for a) maximum PAE asillustrated by curve 200 and b) constant gain as illustrated by curve202 compared with (c) a conventional (prior art) fixed bias amplifier asillustrated by curve 204. FIG. 7 also shows the power probabilitydensity (pdf) function of an OFDM signal such as WiMax as illustrated bycurve 206. The average efficiency over the full pdf can be computed fromthis information. Curve 200 in FIG. 7 is a plot of optimum PAE versusoutput power for the example single stage 1W HFET amplifier.

Curve 202 of FIG. 7 shows the resulting PAE achieved for dual bias foroptimised gain. At high powers, the PAE is very close to the optimumPAE. At low powers it is somewhat less than the optimum, but stillconsiderably greater than the PAE of a conventional (prior art) fixedbias amplifier as represented by curve 204. The corresponding gain curvefor the dual bias amplifier is shown by curve 302 of FIG. 6.

In another alternative example a query may be used in order for thesupply and bias voltage loci for minimum ACPR to be determined. ACPR isdirectly proportional to the magnitude of the instantaneous errorbetween ideal and actual waveforms. Hence, in the absence of memoryeffect ACPR can be minimised by minimising the instantaneous EVM. FIG. 8illustrates, graphically, the definition of instantaneous Error VectorMagnitude. Referring to FIG. 8, the instantaneous EVM is the magnitudeof the Error Vector 502 and is the difference between the Ideal CarrierVector 500 and the measured Carrier Vector 504. It is clear from FIG. 8that the magnitude of the error vector 502 may be directly calculatedfrom the instantaneous magnitude error δr 508 and phase error rθ 506.

It should be noted that the instantaneous EVM is different from the‘system’ EVM often quoted as a waveform quality metric for wirelessstandards. The system EVM is a measure of the carrier error vector atparticular sampling instants (corresponding to the constellation points)and is usually measured after an ideal matched receive filter. Aconsequence of the ‘memory’ introduced by the filter is that there isnot a one-to-one correspondence between the instantaneous power andsystem EVM.

Returning to the dual bias single stage 1W HFET amplifier example, it isuseful to be able to predict the average efficiency of an amplifier whenoperated with a high peak to average power ratio (PAP) signal. Forexample the power statistics of an orthogonal frequency divisionmultiplexing (OFDM) signal such as that used in the 802.16 (WiMax) orthe 802.11a (WLAN) standard may be approximated by a Rayleighdistribution as shown in curve 206 of FIG. 7. Using signal powerprobability density statistics and the measured instantaneous amplifierPAE, the weighted average efficiency of the dual bias amplifier for anymodulation scheme may be computed as a function of average power.

FIG. 9 shows in table form the average efficiency of the examplecontrolled single stage 1W HFET dual bias amplifier optimised for a)maximum PAE and b) constant gain compared with a traditional fixed biasamplifier for various average amplifier output powers. The table in FIG.9 compares the predicted average efficiency of a conventional (priorart) fixed bias amplifier carrying a 802.11a 64 QAM OFDM signal withthat of two dual bias amplifier variants: one optimised for bestefficiency and the other for constant gain. Note that these results areindicative of the efficiency of the RF amplifier alone, and do notaccount for the efficiency of the supply modulator or of any ancillarycircuits. The table of FIG. 9 summarises the curves of FIG. 7.

FIG. 10 shows the AM-AM compression characteristics of the examplecontrolled dual bias single stage 1W HFET amplifier optimized to achievemaximum efficiency for a constant 10 dB gain target. FIG. 11 shows theAM-PM compression characteristics of the example controlled dual biassingle stage 1W HFET amplifier optimised to achieve maximum efficiencyfor a constant 10 dB gain target. The plots of FIGS. 10 and 11 show themeasured variation of phase with input power whilst supply and biasvoltages are fixed at their optimum values for each output power level.

FIG. 10 and FIG. 11 give an indication of the sensitivity of a dual biasamplifier optimised for constant gain to mis-tracking between the inputpower and the ‘target’ supply and bias voltages. The ‘noise’ in theAM-PM curves shown in FIG. 11 is believed to be an artifact of themeasurement system rather than a characteristic of the device. For anyoutput power in FIG. 10 or FIG. 11, the supply and bias voltages are setto their predicted ideal values. The highlighted points on the surfacesshow the expected gain and phase when the amplifier input power isexactly correct. The curves then show the expected gain and phasevariation if the voltages are held at their calculated values, and theinput power is varied. Hence these Figures show the sensitivity toerrors in the input power, or conversely errors in the supply and bias‘set’ voltages. It can be seen that the sensitivity to mistracking isgreater at high output powers than low output powers. This result isintuitively expected, as the amplifier compression is greater at highoutput power.

FIG. 12 shows the predicted constellation of a constant gain optimiseddual bias amplifier, and FIG. 14 shows the predicted spectrum of aconstant gain optimised dual bias amplifier, both carrying a 64 QAM OFDMsignal at 10 dB back-off. FIG. 11 and FIG. 12 are indicative of theperformance expected with a WiMax signal, which has a similar modulationformat. It can be seen that the stringent EVM performance required for ahigh order OFDM signal should in principle be attainable using a ‘selflinearised’ dual bias amplifier.

In the above description, there has been described an example way inwhich the device may be characterised, and the data resulting from suchcharacterisation stored in a database which may then be accessed toobtain preferred or optimum operational characteristics to meet specificor predefined system performance objectives. The characterisation of thedevice in this way and the use of a database in this way is only oneexample of achieving the benefits of the present invention. In analternative a model may be used.

The use of a model rather than a database provides estimates in place ofhaving to search a large database. Updating a model in real-time is lessonerous than updating a database. A model can be tapped into in the sameway that a database can. The model may be created using measurements.The inputs to the model may be from the group comprising bias voltage;supply voltage; input power; input phase; temperature; device periphery;and load impedance. The outputs from the model may be from the groupcomprising: output power; output phase; gain; supply current; adjacentchannel power; error vector magnitude; correlation coefficient.

With reference to FIGS. 2 to 13 it has been described above how anexemplary device may be characterised, and how the measurements obtainedby such characterisation may be used for joint optimisation of supplyand bias voltages for prescribed goals. It is further described theperformance gains that can be thereby obtained. The practicalimplementation of such a technique is now described by way of referenceto particular non-limiting embodiments.

An arrangement in accordance with a first embodiment of the invention isnow described with reference to FIG. 14. Elements of this Figure whichcorrespond to elements shown in FIG. 1 are identified by like referencenumerals. FIG. 14 illustrates an embodiment in which the invention isimplemented in the analogue domain.

In the embodiments of FIG. 14, and subsequent embodiments describedherein, there is shown an amplification stage 100 comprising a single RFamplifier. It should be understood, as discussed hereinabove, that theinvention is not limited to any specific amplifier arrangement.

In the arrangement of FIG. 14 the amplifier 100 receives an RF inputsignal at a first input port on line 141, a continuously variableamplifier supply voltage on a line 108 at a second input port, and anamplifier bias voltage on line 110 at a third input port. The amplifiergenerates an RF output signal on line 142 at an output port.

The RF amplifier is fed with a non-constant envelope signal (e.g. OFDM,CDMA) as an RF input signal on line 141, to which no pre-distortion hasbeen applied. The RF input signal on line 141 is provided as an input tothe amplifier 100 via an optional delay stage 122. As known in the art,the delay stage 122 is optionally provided to delay the input signal tobe applied to the amplifier 100 to allow for processing taking place onsignals derived from such input signal and which also form the basis offurther inputs to the amplifier. The RF input signal is generated by amodulator 128, provided on line 140 to delay stage 122.

The continuously variable amplifier supply voltage on line 109 issupplied by a switched mode DC-DC converter 102. The DC-DC converter 102is controlled by an output from a non-linear mapping entity 104. Theamplifier bias voltage on line 110 is provided by the output of anon-linear mapping element 106.

Each of the non-linear mapping entities 104 and 106 receives as an inputa signal representing the envelope of the RF signal to be amplified.This signal is generated on a line 118 at the output of an envelopedetector 108, which receives as an input the RF signal to be amplifiedon line 120.

The relationship between the instantaneous envelope of the RF input 118and the supply voltage 109 is defined by analogue non-linear mappingelement 104. Similarly, analogue non-linear mapping element 106 is usedto define the relationship between the RF envelope 118 and the amplifierbias voltage 110. In general, the mapping functions of elements 104 and106 are likely to be different from one another, although this is notessential.

An arbitrary non-linear function may be used to describe the non-linearmapping required in blocks 104 and 106. Typically, the required mappingsmay be adequately described by a polynomial expansion of order 3. Thisis discussed hereinabove in relation to FIG. 5.

The non-linear functions may be ‘pre-set’ at manufacture. This isdiscussed in further detail hereinbelow.

Thus the analogue non-linear mapping functions are arranged to providean appropriate functionality such that in response to a current detectedlevel of the envelope signal at the respective inputs, a preferable oroptimum bias or supply voltage is generated at the output. Thisnon-linear mapping functionality may be achieved by an appropriatelyarranged analogue circuit.

An arrangement in accordance with a second embodiment of the inventionis now described with reference to FIG. 15. Elements of this Figurewhich correspond to elements shown in FIGS. 1 and 13 are identified bylike reference numerals. FIG. 15 illustrates an embodiment in which theinvention is illustrated in the digital domain.

In this embodiment, envelope detection is performed in the digitaldomain. Similarly, the non-linear mappings for supply and bias voltagesare also carried out in the digital baseband. One advantage of thisembodiment is the ease of implementing delay elements in the digitaldomain, which makes it easier to obtain precise time alignment betweenthe RF envelope input signal and the amplifier supply and bias voltages.

The amplifier supply voltage on line 109 provided by the output of theDC-DC converter 102 on line 109. The DC-DC converter 102 is controlledby an output on line 112 from a non-linear mapping entity 116. Theamplifier bias voltage on line 110 is provided by the output of anon-linear mapping element 138.

Each of the non-linear mapping entities 116 and 138 receive as an inputa signal representing the envelope of the RF signal to be amplified.This signal is generated on a line at the output of an envelope detector130, which receives as an input the RF signal to be amplified. In FIG.15 a modulator 128 is shown which generates the RF signal to beamplified.

The relationship between the instantaneous envelope of the RF input andthe supply voltage 109 is defined by digital non-linear mapping element116. Similarly, digital non-linear mapping element 138 is used to definethe relationship between the RF envelope and the amplifier bias voltage110. In general, the mapping functions of elements 116 and 138 aredifferent from one another.

The implementation of the non-linear mapping elements 116 and 138 may bein a variety of digital formats. For example they may comprise look-uptables storing the measurement database obtained from characterisationof the device.

In general, whether implemented in the analogue or digital domain, themapping means approximate an ideal supply or bias mapping based on theinput signal, based on the known ideal values from the characterisationof the device. Such mappings are chosen to optimise a specific systemperformance parameter, and the ideal values for a given input signallevel may differ according to the system performance criteria to be met.

In practice, the implementation of any embodiment in accordance with theinvention will be in either the analogue domain as illustrated by theexample of FIG. 14, or the digital domain as illustrated by the exampleof FIG. 15. However in a third embodiment, as illustrated by FIG. 16,the analogue and digital arrangements may be provided in parallel.Elements of this Figure which correspond to elements shown in anyprevious Figure are identified by like reference numerals. FIG. 16illustrates an arrangement in which the invention is implemented ineither the analogue or digital domain.

For the purpose of illustration in FIG. 16 a switch 112 is shown whichselects between the output of the non-linear digital mapping element 116or the output of the non-linear analogue mapping element 104, andprovides the selected output to the DC-DC converter 102. Similarly aswitch 114 is provided which selects either the output of the non-lineardigital mapping element 138 or the output of the non-linear analoguemapping element 106 and provides the selected one as the bias voltage online 110.

In a practical implementation such switches 112 and 114 will not beprovided, and one or other of either an analogue or a digitalimplementation will be provided only. Switches 112 and 114 are shown inFIG. 16 merely to distinguish between the different topologies requiredfor digital and analogue implementations, and to provide a basis forillustrating additional optional aspects of the invention in the contextof both a digital and analogue environment. These additional optionalaspects are now further described with further reference to FIG. 16.

The embodiments described with reference to FIGS. 13 and 14 are ‘openloop’ arrangements, and assume that the performance of the amplifier istime invariant. It will be appreciated that it is impractical to carryout an in depth characterisation of production amplifiers and that therewill be small differences in characteristics between individualamplifiers. For applications requiring very high modulation accuracy,the architecture may be enhanced through incorporation of an adaptivedigital pre-distortion (DPD) block 126 as shown in FIG. 16. Thepre-distortion block is shown connected between the output of themodulator 128 and the input to the delay block 122.

The non-linear functions provided by any of the non-linear mappingelements 116, 138, 104, 106 may be ‘pre-set’ at manufacture as mentionedhereinabove. In an enhanced arrangement they may be updated periodicallyduring operation. Such an enhanced arrangement is also illustrated inFIG. 16.

As further illustrated in FIG. 16, a mapping adaptation element 132 isintroduced. The mapping adaptation element 132 receives an output of adown conversion element 144, the input to which is provided by afeedback connection from the RF output of the RF amplifier 100 on line142. The mapping adaptation element 132 thus receives feedback as to theRF output signal generated by the amplification stage. In an enhancedarrangement, the mapping adaptation element 132 is used to compute newcoefficients for the non-linear mapping blocks 104, 106, 116, 138. Asillustrated in FIG. 16 the mapping adaptation element 132 providesoutputs on a signal line 146 which is labelled “A”. The signals on line146 are provided as inputs to each of the non-linear mapping elements116, 138, 104, 106, to provide updated coefficients therefore. Theadaptation provided by the mapping adaptation element 132 mayadvantageously be provided, in embodiments, for a number of reasons. Theadaptation may allow for tracking of thermal effects; changes inamplified device periphery; change of operational frequency; or simplyalteration of system ‘targets’ (e.g. to achieve minimum EVM rather thanminimum ACPR). The provision of feedback information via the downconversion element 144 is not essential to the operation of the mappingadaptation element 132. The mapping adaptation element 132 may simplyprovide the necessary data in order to achieve new system targets. Theprovision of the feedback information from the output of the RFamplifier is advantageous insofar as it allows for dynamic adaptation ofcoefficients, to take into account effects on the operation of theamplifier stage when in use.

The mapping adaptation block 132 may further receive inputs from atemperature detector 124 to assist in the mapping adaptation process. Inpractice, the down-conversion block may be the receiver chain of atransceiver rather than a dedicated block. Typically, a serial digitalbus 146 is used to communicate updated coefficients to the non-linearmapping blocks 104, 106, 116, 138.

The amplification stage may further incorporate supply voltagemodulation means comprising: a plurality of DC supply voltages and meansfor selecting one DC supply voltage dependent on the output of saidnon-linear supply mapping means; means for determining the error betweensaid selected DC supply voltage and the output of said non-linear supplymapping means; and summing means for adding correction voltage to saidselected DC supply voltage, wherein the amplifier supply voltage issubstantially a replica of the output of said non-linear supply mappingmeans.

The present invention has been described herein by way of reference toparticular exemplary arrangements and embodiments, which arrangementsand embodiments do not limit the scope of the invention. The scope ofprotection afforded by the invention is defined by the appendeddependent claims. One skilled in the art will appreciate variations inrespect of the embodiments of the invention presented herein which fallwithin the scope of the appended claims.

1. A method of controlling at least one amplification stage, comprising:a. selecting a linearity objective for the amplification stage; b. independence on an input signal to said amplification stage, determining acombination of supply input and bias input for the amplification stagein order to meet said linearity objective; and c. in dependence on therebeing more than one combination of supply input and bias input formeeting the linearity objective, selecting the combination thatoptimises a further system performance objective for the amplificationstage.
 2. A method according to claim 1, wherein the further systemperformance objective is one or more of: an efficiency objective; anenvelope signal bandwidth objective; or a robustness to productiontolerance objective.
 3. A method according to claim 1 wherein the supplyinput and the bias input vary in dependence on a variation in theenvelope of the input signal or the power of the input signal and thestep of determining a preferred combination is based on an instantaneousvalue of the input signal.
 4. A method according to claim 1 furthercomprising the steps of: a. measuring at least one amplifier dependentcharacteristic in dependence on at least one amplifier independentcharacteristic; and b. determining a preferred combination of bias andsupply inputs to achieve the specific system performance objective basedon said measurements.
 5. A method according to claim 4 furthercomprising the steps of: a. creating a searchable database of saidamplifier dependent and independent characteristics; b. searching saidmeasurement database to simultaneously determine the optimum combinationof bias and supply voltage at each input power over the measurementrange to achieve specific system performance objectives; c. wherein stepof applying the supply voltage and the bias voltage is based on saiddetermined combinations.
 6. A method according to claim 4 furthercomprising the steps of: a. measuring a plurality of amplifier dependentcharacteristics in dependence on a plurality of amplifier independentcharacteristics; b. creating a model of the amplifier operating foremulation of said measured amplifier characteristics; c. determiningfrom said model the optimum combination of bias and supply voltage ateach input power over the measurement range to achieve specific systemperformance objectives; d. wherein step of applying the supply voltageand the bias voltage is based on said determined combinations.
 7. Amethod according to claim 6 wherein said model is in real-time ornon-real-time.
 8. A method according to claim 4 wherein the plurality ofamplifier independent characteristics are from the group comprising biasvoltage; supply voltage; input power; input phase; temperature; deviceperiphery; and load impedance.
 9. A method according to claim 4 whereinthe plurality of amplifier dependent characteristics are from the groupcomprising: output power; output phase; gain; supply current; adjacentchannel power; error vector magnitude; correlation coefficient.
 10. Amethod according to claim 6 wherein the inputs to the model are from thegroup comprising bias voltage; supply voltage; input power; input phase;temperature; device periphery; and load impedance.
 11. A methodaccording to claim 6 wherein the outputs from the model are from thegroup comprising: output power; output phase; gain; supply current;adjacent channel power; error vector magnitude; correlation coefficient.12. A method according to claim 4 wherein said system performanceobjectives comprise highest power added efficiency; highest drainefficiency; constant gain; constant phase; lowest adjacent channelpower; lowest error vector magnitude; highest correlation coefficient.13. A method according to claim 1 wherein the supply and bias inputs areselected in dependence upon one or more previous input signal values.14. An amplification stage for amplifying an input signal, theamplification stage having a supply voltage input and bias voltageinput, comprising: a. detection means for detecting the input signal tothe amplifier; b. voltage selection means for selecting a supply inputand bias input set for the amplification stage in dependence on thedetected input signal, wherein the selected supply and bias inputs areselected to meet a linearity objective for the amplification stage; andfurther wherein in dependence on there being more than one supply inputand bias input set for meeting the linearity objective, selecting theset that optimises a further system performance objective for theamplification stage.
 15. The amplification stage of claim 14 wherein thefurther system performance objective is one or more of: an efficiencyobjective; an envelope signal bandwidth objective; or a robustness toproduction tolerance objective.
 16. An amplification stage according toclaim 14 wherein the voltage selection means comprises: a. a non-linearmapping element for receiving the detected input signal and generatingthe supply input; and b. a non-linear mapping means for receiving thedetected input signal and generating the bias input.
 17. Anamplification stage according to claim 16 wherein each of the respectivenon-linear mapping means is adapted to approximate an idealised mappingfor the detected input signal to meet the specific system performanceobjective.
 18. An amplification stage according to claim 16 wherein eachof said respective non-linear mapping means is a digital linear mappingmeans.
 19. An amplification stage according to claim 16 wherein each ofsaid respective non-linear mapping means is a digital linear mappingmeans.
 20. An amplification stage according to claim 14 wherein thenon-linear mapping means are configured in accordance with measuredresults for the amplification stage performance.